About This Page.I have become aware that this page was not very well organized. It flowed along in a kind of stream of consciousness style. That may be OK for modern literature but not for a site dedicated to engineering design. It has been reorganized and should flow more logically than before.
Now to the Article.Although some might say the output is the heart of the amplifier, a better analogy would be to say the output tubes are the lungs. That being the case the phase inverter and driver becomes the heart. Even if the output is clean as can be, a poorly designed phase inverter and driver circuit can ruin the sound. Let's examine the requirements for a phase inverter and see how well the various circuits can meet them.
The right way and the wrong way.There are two basic ways to design an amplifier, the right way and the wrong way. The wrong way is not to worry about such minor details as distortion and balance and put on as much feedback as the system will stand without oscillating. Theoretically, the feedback is supposed to clean up the sound.
The right way is to worry about everything. Get the distortion in each stage as low as possible and come up with a really good sounding amplifier even before the application of a single dB of feedback. Then after the application of a modest amount of feedback, say 20 dB, the amplifier will have that sound that makes people's mouths drop open in amazement.
My Design Philosophy.Over the last few months I have come to several conclusions about tube audio amplifier design.
- Balanced drive to the output tubes is very important in order to maintain low distortion.
- There should be only one RC low-frequency roll-off inside the global feedback loop. Failing that there should be as few as possible.
- The fewer stages inside the feedback loop the better.
- Stages ahead of the output tubes and inside the feedback
loop should have as much head room as possible.
Balanced Drive.As you might expect unbalanced drive to the push-pull power output tubes can increase the amount of distortion. Not only that but the amount of distortion needs to be balanced as well. As you may remember some of the phase inverter circuits discussed in Phase Inverters, Phase Splitters, and Drivers deliver more distortion from one output than from the other. You have probably heard that a push-pull output cancels even harmonics. It can only cancel harmonics generated in the output stage itself. If the output receives a signal that is already distorted it can't do a thing to clean it up. Thus, distortion components from the phase inverter are not canceled by the outputs but the higher level is distributed to the other side which equalizes the distortion at the higher level. Low, and nearly equal, levels of distortion from the driver/inverter section of the amplifier are vital.
That eliminates many, but not all, of the phase inverter circuits. The long tail pair showed a lot of promise at first because it did deliver nearly equal and low levels of distortion from the two outputs. Further investigation revealed a flaw in the simpler circuits. One way of applying feedback to such a circuit is to connect it back to the grid of the other tube as shown below.
For a verbal description click here.
The signal which comes back from the output transformer secondary is just a little less in voltage than the input signal and in phase with it. The difference between the two inputs is below the input signal by the same number of dB as the amount of feedback. For example, if you have 20 dB of feedback* on the amplifier and the input signal is 1 volt the difference between the input and fed back signal is 0.1 volts. The fed back signal is 0.9 volts.20 dB is a factor of 10 in voltage. If the value of dB is positive, the voltage is 10 times the reference. If negative, the voltage is 1/10 of the reference. Technically we should say an amplifier has a negative number of dB of feedback. The amplifier's overall gain is reduced by the number of dB of feedback, in this example 20 dB. But the convention is to just express it as a positive number and let it be understood.I have only the two triodes on my breadboard. I can simulate the feed back condition by using a 9/10 voltage divider to apply 9/10 of the input level to the bottom grid in the circuit above. When I do this I find a large imbalance between the two outputs. The reason is the amount of common mode signal that is coming through.
The long tail pair is a differential amplifier. It amplifies the difference between the two grid voltages. Ideally if the exact same signal were applied to both grids the output would be zero. This is called common mode rejection. However it isn't zero because there is a resistor in the cathode (I almost typed emitter) circuit. The resistor is large but it isn't infinite. When identical voltages are applied to the two grids there is some variation of the voltage at the cathodes. This causes alternating current in the cathode resistor which in turn causes AC in each plate resistor. The two voltages are in phase.
The combination of the input signal and fed back signal have a common mode component and a normal mode component. These two add together in the outputs. The common mode component is smaller but not zero. The result is a considerable imbalance in the output levels, as much as 33%. Some designs I have seen employ feedback from a balance point later in the amplifier to the cathodes to cancel out the imbalance signal. I would rather not have the imbalance in the first place instead of adding more parts to balance it.
Of course the feedback can always be taken to the cathode of a single ended triode preceding the long tail pair. This was done in many commercial designs from the 1960s.
Another trick I have seen is to place a 100 ohm resistor in series with the large resistor in the cathodes of the LTP. Feedback is taken to the junction of this 100 ohm resistor and the cathode resistor and also capacitively coupled to the grid of the other triode. (The triode opposite the one that receives signal.) This helps but there is still an imbalance of about 5 percent.
The imbalance can easily be cured by using a constant current sink in the cathode circuit such as a pentode tube or a transistor. More about this later.
Only One RC Low Frequency Roll-off.The output transformer rolls off at low frequencies. When the power output is at, or an appreciable fraction of the maximum power, there is hysteresis distortion. The lissajou pattern becomes a parallelogram. But at low power levels the lissajou pattern remains an ellipse and the amplitude rolls off as an RL (resistance inductance) high-pass filter.
The ideal case would be no RC time constants at all in the feedback loop. I have in fact played with such direct coupled amplifiers. The major problem is maintaining balance in the face of tube drift. A balanced condition could only be maintained for a few minutes before the balance control would have to be readjusted. A chopper circuit would have to be employed to keep it working properly. (Yes, I have worked with vacuum tube analog computers.) The roll-off frequency of the only RC circuit should be much lower than, ideally by a factor of ten, of the roll-off frequency of the output transformer. Classic Bode analysis will allow as many RC time-constants as necessary as long as they don't effect the gain and phase margins. More about this later.
In the simple circuit above although there are two capacitors they are operating on opposite sides of the balanced circuit and act as a single RC roll-off.
There should be only one RC time-constant circuit that effects the low frequency end. Cathode bypass capacitors must be removed or the cathodes grounded. The dropping resistor to a screen grid and the capacitor to ground constitute a low frequency roll-off RC circuit. Screen grid voltages must be regulated by a zener diode or other means.
Minimum Number of Stages in the Feedback Loop.Keeping the stage count low is not just because of KISS, (keep it simple stupid), A problem that plagues all feedback circuits is time delay in the amplifying devices. That time delay combined with the use of too much feedback is what gives transistor amplifiers their harsh sound. The same sound could be achieved with tubes through the use of too many stages and too much feedback. So, the goal is to minimize the number of stages and use feedback in moderation.
Head Room in Early Stages.The output stage should go into overload way before anything else does. The reason is recovery from overloads. When there is negative feedback around an amplifier the feedback is forcing the output wave to match the input. When it doesn't match, the voltage inside the loop changes to compensate. For example, when the output stage goes into overload the feedback will cause the driver stage to produce more output in an attempt to keep the waves matched. This can be seen in the oscilloscope photographs below. The upper wave is the voltage at the grid, inside the feedback loop, of a triode amplifier stage. The lower wave is the voltage at the plate of the same stage.
In the first picture both waves are good sine waves because overload has not started yet. In the second picture the tube is going into cutoff and the feedback is telling the grid voltage to "get more negative" in a vain attempt to make the tube draw less current so the output voltage will go higher. The current is already zero and because the tube only conducts in one direction the current can't get any lower than that.
In the second picture the feedback forces the wave more negative than normal in an attempt to keep the output wave from flat topping. This was a circuit built to illustrate this phenomenon.
When I drove the amplifier harder the bottom of the output wave would flatten but the top of the grid wave would not show a narrow peak because of grid current. However the steady-state test can't show what really happens. In the photo below a burst generator was used to see what happens after a transient overload.
When the tube is driven into grid current the charge on the coupling capacitor changes which changes the bias on the grid for a not-so-short time after the transient. The grid acts like the plate of a diode and rectifies some of the signal and adds the charge to the capacitor. After the transient overload ends the capacitor discharges through the grid resistor and returns to its equilibrium condition.
I know there is at least one engineer out there screaming at the computer screen that it is possible to use more than one RC roll-off as long as the time constant of one is short and all the others are much longer than that. That is the result of classic Bode plot stability analysis. That works fine as long as there is no overloading taking place. When an overload does occur the long time constant capacitors will charge up and then take a long time to discharge. The end result is very poor overload recovery.
Issues in Amplifier Design.The tube manual data for most power tubes states maximum permissible values of the grid resistors. For cathode bias this is usually 470 k ohms and for fixed bias it is 100 k ohms. This is not true for all power tubes and you should look up the maximum value for the particular tube type you are using.
There is no such thing as a perfect vacuum. The vacuum of space in low earth orbit is not nearly as good as that at the distance of the moon's orbit. Even so this low earth orbit vacuum is still considerably better than that within a vacuum tube. Some of the gas molecules become positive ions and are accelerated towards the cathode by the electric field. They are neutralized when they meet up with the space charge surrounding the cathode. Before that happens a few can impact the grid imparting a positive charge to it. A more positive grid will increase the plate current. Increased plate current can increase the number of ions which imparts more positive charge to the grid. If the resistance in the grid circuit is too high this can turn into a runaway condition which can ultimately destroy the tube. The manufacturer specifies a maximum grid resistance to ensure that this runaway condition never gets started.
When cathode bias is used the tube has a built-in regulator. If the grid starts to build up a positive charge the increase in plate current will cause the voltage at the cathode to become more positive. This diminishes the effect of the positive going grid. Because this happens relatively slowly a bypass capacitor will not interfere with this self protective effect. This self protection means that a large grid resistor can be used without the risk of damaging the tube.
Some experimenters are using zener diodes and even light emitting diodes in the cathode circuit to produce bias. In these circuits the bias voltage changes by a very small amount as plate current is changed. When this circuit configuration is used the bias should be treated as fixed bias and the lower value of grid resistance used. In general higher power is available using fixed bias.
The resistance coupled amplifier circuits which have the lowest distortion have relatively high output resistance. For the 12AX7 which delivers high gain and low distortion typically has a 270 k ohm resistor as a plate load and its plate resistance is 80 k ohms. On the other hand is the 12AU7 which delivers lower gain but much higher distortion, typically 4 percent. It can operate with plate loads as low as 22 k ohms and could drive power tube grids with 100 k ohm resistors to AC ground but the high distortion is a concern.
Some of the low distortion circuits I have tested and will show below will not even begin to drive the 100 k ohm grid resistors. Most of them need to look into 470 k or 1 Meg ohm. One possible solution is to drive the output tubes with direct coupled cathode followers. The large resistors and coupling capacitors will be in the grids of these cathode followers. The tube type can be selected so grid current occurs at a much higher level than the grid current in the output tubes themselves. The cathode followers will drive output tube grid current while introducing little distortion. Because the grid current point of the followers is higher it is much less likely that they will be driven into grid current so the capacitors will not get extra charge which will then have to leak off. To deliver low distortion the resistor in the cathode of a cathode follower needs to be returned to a large negative voltage. This adds extra complexity to the power supply which I would rather do without.
The Williamson Circuit.The Williamson amplifier circuit was very popular through the 50s and 60s. It was developed in England and quickly adopted and adapted to American valves, er, tubes. The British version of the 6SN7 valve had slightly different characteristics than the American 6SN7 tube. A few value changes made it work as well as it had in England and the circuit was used in many commercially made amplifiers as well as home brewed projects.
The output stage usually employed tubes that were somewhat higher power than the operating point would require, such as 807s in a 25 watt amplifier. It was felt that this gave lower distortion than for a pair of tubes pushed to their limit. I have found this to be true.
The sections that made a Williamson amplifier, a Williamson amplifier, were the split load phase inverter and driver. The circuit is given below.
Figure 1 Original Williamson Inverter/Driver.
For a verbal description click here.
The two RC networks at the input are to keep infrasonic frequencies from entering the amplifier. These inaudible signals, mostly from turntable rumble, eat up amplifier headroom and can lead to audible distortion.
The circuit will look familiar but different. The plate of the first amplifier is directly coupled to the grid of the split load phase inverter rather than through a capacitor. Although this saves one capacitor and two resistors, reducing component count is not the purpose of this modification. It is to eliminate one RC time constant from the signal path. The fewer there are the better global feedback will behave. In the circuits above the grid is already at a positive potential and of course so is the plate. Why not make them the same and eliminate the capacitor.
The phase splitter is followed by a push-pull triode amplifier. Those familiar with the Williamson circuit will see that I have added a modification to it. Actually I have modified someone else's modification. They had placed the balance control in the plate load of one of the triode amplifiers. A practical potentiometer has quite a lot of parasitic capacitance. Adding this extra capacitance in only one plate could cause imbalance at high frequencies even if balanced at low and mid. Moving the balance control to the cathode circuit permits use of a lower resistance pot which moves the roll off beyond the audible range. The pot in the cathode circuit introduces a small amount of degeneration and close to equal amounts of stray capacitance in both sides of the driver.
The performance is as shown in the diagram. The voltage gain is measured from input to one of the outputs. The justification for using 17 volts and 25 volts as the test levels is this. Driving a pair of 6L6s to 26 watts output requires 16 volts RMS per grid. A pair of 6550s driven to 55 watts requires 17 volts RMS per grid. A pair of 6CA7/EL34s requires 25 volts RMS per grid for 54 watts output.
Most of the distortion is taking place in the driver. With the output still set to 25 volts and measuring distortion at the output of the phase splitter it was 0.16% at the plate and 0.14% at the cathode.
Suppose you are going to build a Williamson amplifier using EL34s. The tube manual data for a 6CA7/EL34 gives 1.6% distortion at 54 watts output. Distortion does not add by simple addition but as the square root of the sum of the squares. Thus the overall distortion would be 1.9 squared + 1.6 squared equals 6.17 and the square root of that is 2.48%. 20 dB of global feedback would reduce that to a respectable 0.248%.
Now here's a real kicker! If you leave out the second 6SN7 and drive the output tubes right from the phase splitter the distortion figures are as follows.
For 17 volts, THD = 0.9% at both cathode and plate.
For 25 volts, THD = 1.3%.
With 100 k ohm resistors on the other side of the coupling caps instead of the 470 k the distortion is,
For 17 volts, THD = 0.86%.
For 25 volts, THD = 1.25%.
YOU CAN LEAVE OUT THE DRIVER AND GET LOWER DISTORTION. Of course there is not as much overall gain for global feedback. It may be necessary to add a stage before the 6SN7 to bring the gain up to a sufficient level. This stage should not contribute much distortion as it will be operating at a very low level.
The total distortion for the first part of a Williamson circuit and a pair of 6CA7/EL34s becomes 2.03%.
Redesigning the Williamson.The above circuit is 50 years old if it's a day. I decided to try to reduce the distortion by using somewhat more modern tubes. The 6SN7 goes back to the world war II years.
I used a 12AX7 for the first amplifier. This tube performs best with a relatively high plate voltage while the direct coupling to the phase splitter wants the plate voltage to be somewhat low, ideally 1/4 of the B+ voltage.
For maximum output voltage swing the split load phase inverter needs 1/4 of Ebb (B+) across the cathode resistor, 1/2 of Ebb across the tube, and 1/4 of Ebb across the plate resistor. As the tube current decreases the cathode voltage goes down by 1/4 Ebb and the plate voltage goes up by 1/4 Ebb. As the tube current increases the cathode voltage goes up by 1/4 Ebb and the plate voltage goes down by 1/4 Ebb. They meet in the middle at 1/2 Ebb. This assumes an ideal amplifying device. Since tubes are far from ideal it is unrealistic to expect a peak-to-peak output voltage of 1/2 Ebb. The grid voltage of the inverter, and hence the cathode voltage should be somewhere in the neighborhood of 1/4 Ebb. If Ebb = 400 volts then the grid voltage of the inverter and the plate voltage of the amplifier should be about 100 volts. Let's examine how the distortion of a 12AX7 changes with the value of its plate voltage.
12AX7 Williamson First stage alone.
Vo = 10 v RMS. Ebb adjusted to achieve desired Eb.
Rk = 4700
Rb = 240 k
Rk = 1000
Rb = 270 k
Rk = 1000
Rb = 100 k
80 2.9 2.55 6.0 100 1.75 1.38 2.80 120 1.2 0.86 1.5 140 1.05 -- 1.0 160 0.96 -- -- 180 0.86 -- -- 200 0.78 -- --
As the above makes clear a 12AX7 is not happy with a low plate voltage. We can cheat a little and push up the voltage to about 120 volts but power dissipation in the cathode and plate resistors limits just how far we can go with this.
Figure 2 Improved Williamson Inverter/Driver Circuit.
For a verbal description click here.
What the Heck is a 12DW7?It's a duo triode in which one is like a 12AX7 and the other is like a 12AU7. I am told it was designed for the jukebox industry which only used one amplifier and needed the tube for the amplifier and phase inverter. In fact the functional description in my Sylvania tube manual gives it as "Amplifier and phase inverter".
Since most of us are building stereo amplifiers we can use a 12AX7 for the amplifier tubes in both channels and a 12AU7 as the phase inverter in both channels. If you want to build monoblocks, you will just have to spring for the 12DW7/7247 which is available from AES.
The circuit.It's almost identical to the older Williamson circuit. Self balancing cathode coupling has been sacrificed to allow cathode degeneration to reduce distortion. More cathode feedback has been introduced than would be allowed by the resistance necessary for the proper bias. High gain is not required in this stage. Most of the gain comes from the first 12AX7-like triode.
Cc and Rc are frequency compensating components. They need to be adjusted to match the output transformer which is being used. They prevent oscillation or ringing at high frequencies caused by phase shifts in the output transformer at high frequencies. Typical values are 15 k ohm and 82 pf. Start with these values, drive the amplifier with a square-wave and adjust the values up or down to minimize or eliminate ringing.
Like the old Williamson this new one can do without the 12BH7 driver. Rough measurements gave, at 17 volts the distortion directly from the split load phase inverter is 0.6% and at 25 volts it is 0.9%. Only slightly higher. The gain under these conditions is 52. These numbers make it worth a figure.
Figure 3 Improved Williamson Inverter Without the Driver Circuit.
For a verbal description click here.
Now we are getting somewhere. The gain is high enough to give a sensitivity of 3.18 volts for 26 watts from a pair of 6L6s with 20 dB of feedback, there is only one RC low frequency roll-off inside the feedback loop, and distortion is low enough to have little significance compared to the distortion in the output tubes. Although I haven't built an amplifier using this circuit I think I will sometime in the future.
The Long Tail Pair.A phase inverter circuit often used in high fi and guitar amplifiers in the 1950s and 1960s is the long tail pair. This is in current terminology a differential amplifier. The one shown below is a direct coupled version and is a variation of one used in a commercial amplifier that had cathode loaded output stages. It had to be capable of delivering 50 volts RMS per grid at a reasonable distortion level.
Figure 4 Direct Coupled Long Tail Pair Phase Inverter.
For a verbal description click here.
This is not a split load phase inverter but the original phase inverter which obtained the inversion by passing the signal through one more stage. This version overcomes the common mode rejection problem by applying feedback to a single ended stage before the inverter part of the circuit.
Note that the asymmetry of distortion has diminished from about 7 to 1 to about 2 to 1. In spite of the remaining asymmetry it was used by many amplifier manufacturers, particularly Eico, in some amplifiers which sounded very good. Perhaps unbalanced distortion is not as important as I believe it to be. However, I still am not comfortable with this condition.
The DC plate voltage on the first 12BH7 is applied directly to the grid of the upper triode in the LTP (long tail pair) and to the grid of the lower triode through a low pass filter consisting of the 1 Megohm resistor and 0.22 uf capacitor. This places both grids at the same DC potential while applying signal only to the grid of the upper triode. The 15 k ohm resistor in the cathodes sets the plate current to the correct value for proper operation.
The AC signal which is applied to the grid of the upper triode causes a variation of plate current as in a normal amplifier. This varying current, which is in phase with the grid voltage, flows through the cathode resistor as well as the plate resistor. The varying current in the cathode resistor causes a voltage variation which is in phase with the grid signal. This voltage is directly coupled to the cathode of the lower triode in the LTP. The grid of the lower LTP triode is grounded for AC through the 0.22 uf capacitor. This makes it a grounded grid amplifier.
In normal Class A operation the grid has a negative DC bias on it. In this case the grids are at about 90 volts while the cathodes are at about 100 volts. When the AC signal drives the cathode of the lower triode more positive the grid and cathode are becoming farther apart. This is an increase of the bias which results in a decrease of plate current. We see that the plate current in the lower triode is out of phase with the plate current of the upper triode. If they were exactly equal they would completely cancel and there would be no voltage variation at the cathode to change the bias on the lower triode and there would be no plate current variations. Oops? This contradiction means that the plate current variations in the lower triode are not as large as those in the upper triode. To have the same output voltage from both triodes in the LTP the load resistor of the lower triode must be slightly larger than the load resistor for the upper one.
Lets look at it from the standpoint of current. The upper tube is producing cathode current. Some of this alternating current flows to ground through the 15 k ohm resistor and the rest enters the cathode of the lower tube. If you write the equation IR + IV2 = IV1 you see that the currents in the two tubes are 180 degrees out of phase. If there were a device that had infinite impedance for AC while still passing DC the alternating currents in the two tubes would be equal and the load resistors could be made equal. The larger the common cathode resistor, the better the balance.
This version also contains two low frequency roll-offs. The two 0.22 uf caps looking into 100 k ohm resistors make up one and the other is the 1 Meg and 0.22 in the grid of the lower tube. As this network rolls off the same signal will begin to be applied to both grids. Although there will still be a considerable amount of in-phase signals applied to the two grids of the outputs, they will be cancelled in the output transformer. This causes a low frequency roll-off and the accompanying phase shift inside the feedback loop. It is possible to design a long tail pair that will do better than this.
Here's the circuit I came up with.
Figure 5 Long Tail Pair with Transistor Current Sink.
For a verbal description click here.
No, I haven't lost my mind. Anything I might say to defend my use of a transistor is going to sound like rationalization. It's there, learn to live with it. It makes a very constant current sink. This is the device mentioned above that has infinite AC impedance while passing DC current. It yields a circuit with a very high common mode rejection ratio, although not infinite, and low distortion.
Here is how it works. Lets say the collector current tries to increase. The emitter current will also try to increase which makes the drop across the 1.6 k ohm resistor increase. Because the base voltage is held very constant by the voltage divider of the 3 k and 1.1 k ohm resistors the bias on the base-emitter junction will be decreased. This will decrease the base current and cancel out most of the collector current increase. The collector current has to change by a small amount to make the base current change, but the change is very small. The collector current is 3 mA and most 3904s have a current gain of more than 100 so the base current is about, or less than, 30 microamps. The divider current is 4.88 mA. The base current is 0.615% of the divider current.
As you can see the distortion was very low, much lower than I had expected. Also the distortion at both outputs was very similar. When feedback was simulated the balance remained perfect. The distortion changed though. The plate on the left gave 0.09% while the right hand plate gave 0.175%. This low level of distortion means the distortion imbalance can be disregarded. The transistor can have an overload problem. My first design only used a 12 volt negative power supply and I had to re-engineer it for a -20 volt supply. If I were going to use this circuit I think I would go up to a -30 volt supply.
Now Let's Do It With a Tube.
Figure 6 Long Tail Pair with Pentode Current Sink.
For a verbal description click here.
I selected a pentode rather than a triode because a pentode by its nature is a current sink and the control grid will not have to do as much to make it into one. The 6BH6 was the best tube in this application. The resistor in the cathode works exactly like the resistor in the emitter circuit above to regulate the current and hold it constant. The -70 volt supply MUST be regulated with zener diodes for low noise and stable performance. I often use two or three diodes in series to make up the exact voltage I need for a particular purpose.
As you can see the distortion figures are much lower than would ever be needed. When feedback was simulated the left hand plate gave 0.15%, and the right hand plate 0.19% distortion. This looks like the one I would recommend and use myself.
It has one major drawback. It will not drive 100 k ohm grid resistors in the following stage. If fixed bias is to be used in the output tubes they must be driven by direct coupled cathode followers. That is an additional complication along with the negative regulated power supply. If one wants to build the highest quality amplifier with no compromises this circuit might be used along with regulated screen grid voltages as in the 6L6 monoblock amplifier. This phase inverter would, without a doubt, give a good account of itself in a cathode biased amplifier although the negative power supply would still be necessary for the negative voltage for the cathode.
Traditionally Applied Feedback.Negative feedback in audio amplifiers is all but universally applied to the cathode of the first stage after the last user adjusted control, usually the volume or channel balance control. The Williamson circuit is ready made for this connection and there is plenty of gain. The direct coupled long tail pair also has an available cathode for the application of feedback. The gain of 50 is a little on the marginal side but is still enough. However, the current sink long tail pair is a little low.
Now We Need a Preamplifier.If you run Figure 6 through the feedback program using 20 dB of feedback, 6L6 outputs, and an 8 ohm speaker, you find that it needs 6.685 volts of input to drive it to full power. That's a bit much and so it would be a good idea to use an amplifier outside the feedback loop to bring up the input. If you want 1 volt to drive to full power then you need a stage with a gain of 6.685. If you want to use the classic line level of 1 milliwatt in a 600 ohm load (0.7746 volts) You would need a gain of 8.63. Here is an amplifier stage that will give a fairly wide range of gain at very low levels of distortion.
You may be wondering why I have gone for such low distortion while I didn't indicate it was that important above. This amplifier is NOT meant to be inside the global feedback loop. As such it needs to have an inherently low distortion level because nothing is going to improve it later.
Happy building and good luck. Keep watching, there will be more in the future for the builder or the curious experimenter.
Next; An Amplifier Test Bed.
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This site begun March 14, 2001
This page last updated December 3, 2010.